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AD8361 Datasheet(Fiches technique) 9 Page - Analog Devices

Numéro de pièce AD8361
Description  LF to 2.5 GHz TruPwr™ Detector
Télécharger  16 Pages
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Fabricant  AD [Analog Devices]
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AD8361 Datasheet(HTML) 9 Page - Analog Devices

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The AD8361 is an rms-responding (mean power) detector pro-
viding an approach to the exact measurement of RF power that
is basically independent of waveform. It achieves this function
through the use of a proprietary technique in which the outputs
of two identical squaring cells are balanced by the action of a
high-gain error amplifier.
The signal to be measured is applied to the input of the first
squaring cell, which presents a nominal (LF) resistance of 225
between the pin RFIN and COMM (connected to the ground
plane). Since the input pin is at a bias voltage of about 0.8 V
above ground, a coupling capacitor is required. By making this
an external component, the measurement range may be extended
to arbitrarily low frequencies.
The AD8361 responds to the voltage, VIN, at its input, by squaring
this voltage to generate a current proportional to VIN squared.
This is applied to an internal load resistor, across which is con-
nected a capacitor. These form a low-pass filter, which extracts
the mean of VIN squared. Although essentially voltage-responding,
the associated input impedance calibrates this port in terms of
equivalent power. Thus 1 mW corresponds to a voltage input of
447 mV rms. In the Application section it is shown how to match
this input to 50
The voltage across the low-pass filter, whose frequency may
be arbitrarily low, is applied to one input of an error-sensing
amplifier. A second identical voltage-squaring cell is used to
close a negative feedback loop around this error amplifier.
This second cell is driven by a fraction of the quasi-dc output
voltage of the AD8361. When the voltage at the input of the
second squaring cell is equal to the rms value of VIN, the loop
is in a stable state, and the output then represents the rms value of
the input. The feedback ratio is nominally 0.133, making the
rms-dc conversion gain
×7.5, that is
VOUT = 7.5
× VIN rms
By completing the feedback path through a second squaring cell,
identical to the one receiving the signal to be measured, several
benefits arise. First, scaling effects in these cells cancel; thus, the
overall calibration may be accurate, even though the open-loop
response of the squaring cells taken separately need not be.
Note that in implementing rms-dc conversion, no reference
voltage enters into the closed-loop scaling. Second, the tracking
in the responses of the dual cells remains very close over tempera-
ture, leading to excellent stability of calibration.
The squaring cells have very wide bandwidth with an intrinsic
response from dc to microwave. However, the dynamic range
of such a system is fairly small, due in part to the much larger
dynamic range at the output of the squaring cells. There are
practical limitations to the accuracy with which very small error
signals can be sensed at the bottom end of the dynamic range,
arising from small random offsets; these set the limit to the
attainable accuracy at small inputs.
On the other hand, the squaring cells in the AD8361 have
a “Class-AB” aspect; the peak input is not limited by their
quiescent bias condition, but is determined mainly by the
eventual loss of square-law conformance. Consequently, the top
end of their response range occurs at a fairly large input level
(about 700 mV rms) while preserving a reasonably accurate
square-law response. The maximum usable range is, in practice,
limited by the output swing. The rail-to-rail output stage can
swing from a few millivolts above ground to less than 100 mV
below the supply. An example of the output induced limit: given
a gain of 7.5 and assuming a maximum output of 2.9 V with a 3 V
supply; the maximum input is (2.9 V rms)/7.5 or 390 mV rms.
An important aspect of rms-dc conversion is the need for
averaging (the function is root-MEAN-square). For complex RF
waveforms such as occur in CDMA, the filtering provided by
the on-chip low-pass filter, while satisfactory for CW signals above
100 MHz, will be inadequate when the signal has modulation
components that extend down into the kilohertz region. For this
reason, the FLTR pin is provided: a capacitor attached between
this pin and VPOS can extend the averaging time to very low
An offset voltage can be added to the output (when using the
micro_SOIC version) to allow the use of A/D converters whose
range does not extend down to ground. However, accuracy at
the low end will be degraded because of the inherent error in this
added voltage. This requires that the pin IREF (internal reference)
should be tied to VPOS and SREF (supply reference) to ground.
In the IREF mode, the intercept is generated by an internal
reference cell, and is a fixed 350 mV, independent of the supply
voltage. To enable this intercept, IREF should be open-circuited,
and SREF should be grounded.
In the SREF mode, the voltage is provided by the supply. To
implement this mode, tie IREF to VPOS and SREF to VPOS. The
offset is then proportional to the supply voltage, and is 400 mV
for a 3 V supply and 667 mV for a 5 V supply.
Basic Connections
Figures 32, 33, and 34 show the basic connections for the
micro_SOIC version AD8361 in its three operating modes. In all
modes, the device is powered by a single supply of between 2.7 V
and 5.5 V. The VPOS pin is decoupled using 100 pF and 0.01
capacitors. The quiescent current of 1.1 mA in operating mode
can be reduced to 1
µA by pulling the PWDN pin up to VPOS.
A 75
Ω external shunt resistance combines with the ac-coupled
input to give an overall broadband input impedance near 50
Note that the coupling capacitor must be placed between the in-
put and the shunt impedance. Input impedance and input coupling
are discussed in more detail below.
The input coupling capacitor combines with the internal input
resistance (Figure 13) to give a high-pass corner frequency
given by the equation

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