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MC33067P Fiches technique(PDF) 10 Page - ON Semiconductor

No de pièce MC33067P
Description  HIGH PERFORMANCE ZERO VOLTAGE SWITCH RESONANT MODE CONTROLLERS
Download  16 Pages
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Fabricant  ONSEMI [ON Semiconductor]
Site Internet  http://www.onsemi.com
Logo ONSEMI - ON Semiconductor

MC33067P Fiches technique(HTML) 10 Page - ON Semiconductor

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MC34067 MC33067
10
MOTOROLA ANALOG IC DEVICE DATA
APPLICATIONS INFORMATION
The MC34067 is specifically designed for zero voltage
switching (ZVS) quasi–resonant converter (QRC)
applications. The IC is optimized for double–ended push–pull
or bridge type converters operating in continuous conduction
mode. Operation of this type of ZVS with resonant properties
is similar to standard push–pull or bridge circuits in that the
energy is transferred during the transistor on–time. The
difference is that a series resonant tank is usually introduced
to shape the voltage across the power transistor prior to
turn–on. The resonant tank in this topology is not used to
deliver energy to the output as is the case with zero current
switch topologies. When the power transistor is enabled the
voltage across it should already be zero, yielding minimal
switching loss. Figure 19 shows a timing diagram for a
half–bridge ZVS QRC. An application circuit is shown in
Figure 20. The circuit built is a dc to dc half–bridge converter
delivering 75 W to the output from a 48 V source.
When building a zero voltage switch (ZVS) circuit, the
objective is to waveshape the power transistor’s voltage
waveform so that the voltage across the transistor is zero
when the device is turned on. The purpose of the control IC is
to allow a resonant tank to waveshape the voltage across the
power transistor while still maintaining regulation. This is
accomplished by maintaining a fixed deadtime and by
varying the frequency; thus the effective duty cycle is
changed.
Primary side resonance can be used with ZVS circuits. In
the application circuit, the elements that make the resonant
tank are the primary leakage inductance of the transformer
(LL) and the average output capacitance (COSS) of a power
MOSFET (CR). The desired resonant frequency for the
application circuit is calculated by Equation 6:
L L 2CR
(6)
1
=
π
2
ƒr
In the application circuit, the operating voltage is low and
the value of COSS versus Drain Voltage is known. Because
the COSS of a MOSFET changes with drain voltage, the value
of the CR is approximated as the average COSS of the
MOSFET. For the application circuit the average COSS can be
calculated by Equation 7:
measured at
(7)
CR
1
2
2 * COSS
=
in
V
The MOSFET chosen fixes CR and that LL is adjusted to
achieve the desired resonant frequency.
However, the desired resonant frequency is less critical
than the leakage inductance. Figure 19 shows the primary
current ramping toward its peak value during the resonant
transition.
During this time, there is circulating current
flowing through the secondary inductance, which effectively
makes the primary inductance appear shorted. Therefore,
the current through the primary will ramp to its peak value at
a rate controlled by the leakage inductance and the applied
voltage. Energy is not transferred to the secondary during
this stage, because the primary current has not overcome the
circulating current in the secondary. The larger the leakage
inductance, the longer it takes for the primary current to slew.
The practical effect of this is to lower the duty cycle, thus
reducing the operating range.
The maximum duty cycle is controlled by the leakage
inductance, not by the MC34067. The One–Shot in the
MC34067 only assures that the power switch is turned on
under a zero voltage condition. Adjust the one–shot period so
that the output switch is activated while the primary current is
slewing but before the current changes polarity. The resonant
stage should then be designed to be as long as the time for
the primary current to go to zero amps.


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