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SA5212AFE Fiches technique(PDF) 11 Page - NXP Semiconductors |
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SA5212AFE Fiches technique(HTML) 11 Page - NXP Semiconductors |
11 / 20 page Philips Semiconductors Product specification SA5212A Transimpedance amplifier (140MHz) 1998 Oct 07 11 THEORY OF OPERATION Transimpedance amplifiers have been widely used as the preamplifier in fiber-optic receivers. The SA5212A is a wide bandwidth (typically 140MHz) transimpedance amplifier designed primarily for input currents requiring a large dynamic range, such as those produced by a laser diode. The maximum input current before output stage clipping occurs at typically 240 µA. The SA5212A is a bipolar transimpedance amplifier which is current driven at the input and generates a differential voltage signal at the outputs. The forward transfer function is therefore a ratio of the differential output voltage to a given input current with the dimensions of ohms. The main feature of this amplifier is a wideband, low-noise input stage which is desensitized to photodiode capacitance variations. When connected to a photodiode of a few picoFarads, the frequency response will not be degraded significantly. Except for the input stage, the entire signal path is differential to provide improved power-supply rejection and ease of interface to ECL type circuitry. A block diagram of the circuit is shown in Figure 10. The input stage (A1) employs shunt-series feedback to stabilize the current gain of the amplifier. The transresistance of the amplifier from the current source to the emitter of Q3 is approximately the value of the feedback resistor, RF=7kΩ. The gain from the second stage (A2) and emitter followers (A3 and A4) is about two. Therefore, the differential transresistance of the entire amplifier, RT is R T + V OUT(diff) I IN + 2R F + 2(7.2K) + 14.4kW The single-ended transresistance of the amplifier is typically 7.2k Ω. The simplified schematic in Figure 11 shows how an input current is converted to a differential output voltage. The amplifier has a single input for current which is referenced to Ground 1. An input current from a laser diode, for example, will be converted into a voltage by the feedback resistor RF. The transistor Q1 provides most of the open loop gain of the circuit, AVOL≈70. The emitter follower Q2 minimizes loading on Q1. The transistor Q4, resistor R7, and VB1 provide level shifting and interface with the Q15 – Q16 differential pair of the second stage which is biased with an internal reference, VB2. The differential outputs are derived from emitter followers Q11 – Q12 which are biased by constant current sources. The collectors of Q11 – Q12 are bonded to an external pin, VCC2, in order to reduce the feedback to the input stage. The output impedance is about 17 Ω single-ended. For ease of performance evaluation, a 33 Ω resistor is used in series with each output to match to a 50 Ω test system. BANDWIDTH CALCULATIONS The input stage, shown in Figure 12, employs shunt-series feedback to stabilize the current gain of the amplifier. A simplified analysis can determine the performance of the amplifier. The equivalent input capacitance, CIN, in parallel with the source, IS, is approximately 7.5pF, assuming that CS=0 where CS is the external source capacitance. Since the input is driven by a current source the input must have a low input resistance. The input resistance, RIN, is the ratio of the incremental input voltage, VIN, to the corresponding input current, IIN and can be calculated as: R IN + V IN I IN + R F 1 ) A VOL + 7.2K 70 + 103W More exact calculations would yield a higher value of 110 Ω. Thus CIN and RIN will form the dominant pole of the entire amplifier; f*3dB + 1 2 p R IN CIN Assuming typical values for RF = 7.2kΩ, RIN = 110Ω, CIN = 10pF f*3dB + 1 2 p (110) 10 @ 10*12 + 145MHz The operating point of Q1, Figure 2, has been optimized for the lowest current noise without introducing a second dominant pole in the pass-band. All poles associated with subsequent stages have been kept at sufficiently high enough frequencies to yield an overall single pole response. Although wider bandwidths have been achieved by using a cascade input stage configuration, the present solution has the advantage of a very uniform, highly desensitized frequency response because the Miller effect dominates over the external photodiode and stray capacitances. For example, assuming a source capacitance of 1pF, input stage voltage gain of 70, RIN = 60 Ω then the total input capacitance, CIN = (1+7.5) pF which will lead to only a 12% bandwidth reduction. INPUT OUTPUT + OUTPUT – A1 A2 A3 A4 RF SD00327 Figure 10. SA5212A – Block Diagram NOISE Most of the currently installed fiber-optic systems use non-coherent transmission and detect incident optical power. Therefore, receiver noise performance becomes very important. The input stage achieves a low input referred noise current (spectral density) of 3.5pA/ √Hz. The transresistance configuration assures that the external high value bias resistors often required for photodiode biasing will not contribute to the total noise system noise. The equivalent input RMS noise current is strongly determined by the quiescent current of Q1, the feedback resistor RF, and the bandwidth; however, it is not dependent upon the internal Miller-capacitance. The measured wideband noise was 52nA RMS in a 200MHz bandwidth. |
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